Many applications and implementations that employ integrated switches, such as integrated switched mode power converters, may require that the current flowing through the switches is monitored or compared with given reference values. The reasons for this are related either to regulation or control aspects or to the implementation of additionally required features, such as many kinds of current limitation circuits.
The adopted solutions suitable for silicon integration found so far are always based on the combination of the real switch device carrying the current of interest with a smaller replica or replicas of the switch itself. Typically the real switch device is an integrated MOS transistor through which the load current flows. The switch device and its replica, which can be the unity cell of the larger switch device, are arranged together in a way that a current flows through the replica which is in principle proportional to the interested current. The factor between the current through the switch device and the current through the replica is in principle the integer area scaling factor between the two switch devices. By employing this principle the prior art solutions often do not overcome either the constraint of not having a negative supply rail or the problem that the current flowing in the larger device might have the opposite direction than the current flowing in the replica or in a part of the replicas. This is especially the case in buck converters where the current in the switches always flows towards the external coil. Various conventional current sensing circuits used for switch devices are shown in FIG. 1 to FIG. 3.
In FIG. 1 MOS transistors M1 and M2 are shown acting as a switch device and its replica, respectively. A current Iload flows through the transistor M1, whereas a scaled current Iload/N flows through the transistor M2. Transistors M3, M4, M5 and M6 form a comparator that compares the scaled current Iload/N with a reference current Iref supplied by a current source.
If the current Iload and the scaled current Iload/N enter the drains of the transistors M1 and M2, respectively, the output voltage Vout changes its state when Iload/N=Iref*R1/Rsense (inaccuracies have been neglected) meaning Vout=0 if Iload/N>Iref*R1/Rsense and Vout=VDD if Iload/N<Iref*R1/Rsense. If, however, the current Iload has the opposite direction, which for example may happen in buck converters, the circuit arrangement 1 as shown in FIG. 1 is no longer able to compare the scaled current Iload/N with the reference current Iref. In order to make the circuit work properly, a parallel comparator, which is similar to one formed by the transistors M3 to M6 but mirrored in order to use a negative supply rail, would be necessary. Apart from having additional circuitry the problem is that often no negative rail is available. Moreover, matching issues involving both transistors and resistors may occur and the sensing resistor Rsense introduces a voltage difference in the gate voltage that results in current measurement inaccuracies.
In FIG. 2 a more accurate solution is shown compensating the voltage drop across the sensing resistor Rsense in case that the ON resistance of the transistor M2 is not low enough in comparison with the sensing resistance Rsense. In the circuit arrangement 2 a transistor M3 and a resistor R1 form an additional replica of the switch device M1 and are used as compensating elements. In this case an operational amplifier OPA used as a comparator changes its state if Iload/N>Iref.
However, the current sensing circuit shown in FIG. 2 does not work if the current Iload flows in the opposite direction because the current flowing through the transistor M3 would not notice direction change. Furthermore, matching issues related to transistors and resistors may also appear in the current sensing circuit of FIG. 2.
In FIG. 3 a circuit arrangement 3 is shown that overcomes some of the problems addressed above. The virtual ground created by an operational amplifier OPA is used to read the scaled current Iload/N and send it through the transistor M3. In this case the current Iload can be ejected from the drain of the transistor M1 and the output voltage Vout changes to ground potential if Iload/N>Iref. Nevertheless, this solution requires a fast amplifier, at least in the case of relatively high switching frequencies, and might therefore conflict with the current consumption budget. Moreover, the current sensing circuit shown in FIG. 3 does not work if the current Iload feeds the drain of the transistor M1.
Adapting the circuit arrangement 3 for this case again involves quite complex additional circuitry: typically a fixed current biasing the transistor M3 would be implemented so that the transistor M3 is always conducting even if the current Iload flows in the opposite direction. This might again conflict with minimum current consumption requirements.
Further current sensing circuits are disclosed in the U.S. patent application publications nos. US 2003/0218455 A1, US 2004/0155662 A1, US 2004/0227539 A1 and US 2005/0127888 A1 as well as in the German laid open documents DE 102 58 766 A1 and DE 103 14 842 A1.